A Highly Isolated Co-Polarized Co-Located Electric and Magnetic Dipoles and Its MIMO Performance

Xingning Jia, Dazhi Piao＊

Department of Communication Engineering, Communication University of China, Beijing 100024, China

I. INTRODUCTION

With the unstoppable progress of wireless communication, MIMO technique has been paid incremental attention, because which can increase the channel capacity with linear increase of antenna elements. There are several ways to obtain the diversity required by a MIMO system, such as spatial, polarization and pattern diversity [1]. Compared to the spatial diversity, polarization and pattern diversity can make the MIMO antenna array more compact. Furthermore, by polarization diversity, the vector nature of electromagnetic(EM) wave can be utilized more effectively.It has been suggested that each of the six distinguishable electric and magnetic field polarization components can offer an independent sub-channel in rich multipath environment[2]. Moreover, it has been verified that 1.6-1.7 times of capacity gain (CG) over a single antenna can be obtained by the co-polarized,co-located electric and magnetic dipoles(CCEMD) in an indoor laboratory [3]. Some multi-polarized antennas have been realized,including the design based on loop antenna[4, 5, 8], patch antenna [6] and slot radiator[7], and some analysis about the channel characteristic of polarization diversity have been conducted [8-10], however, the design of the CCEMD based dual-port MIMO antenna is still so few. Moreover, its MIMO channel performance has not been fully investigated,which is of great importance to understanding of the inherent property of the MIMO system using all the six EM components.

The CCEMD antenna presented in [3] is realized separately by a Kandoian loop and a half-wavelength electric dipole with a balun vertical to the plane containing the antenna,which is not a planar structure and not convenient for mass production. Thus, the CCEMD constructed by planar structure with compactness and low pro file is highly desired. In [7],a compact CCEMD antenna has been realized by a metamaterial-based dual-mode loop antenna using a hybrid feeding network, with the isolation of 25 dB between two ports at frequency band around 2.4 GHz, however, the common -10 dB impedance bandwidth of the two ports is only 120 MHz.

In this paper, a compact CCEMD antenna with very high isolation realized by the segmented line based zero-shift-phase-line(ZPSL) loop is presented. The ZPSL were utilized to design electric large antenna having the current along the loop approximately unchanged in phase and flowing in a single direction [11]. Two polarizations are realized by two modes separately excited in the loop,the first one produces an radiation pattern of a two-monopole array, realized by a T-shaped microstrip feed line, and the other mode has a radiation pattern of a magnetic dipole, realized by direct feeding on the loop without any impedance matching construction. By this structure, a very low mutual coupling between the two ports can be obtained, with the simulated value lower than -58 dB around 2.4 GHz. Furthermore, compared with [7], this antenna has a wider impedance bandwidth. A prototype of the antenna is fabricated and tested, the measured common -10 dB impedance bandwidth of the two ports is 280 MHz and the isolation is better than 41 dB over this band. Moreover,the MIMO channel performance of this CCEMD antenna is measured over a channel containing two parallel metal planes, which is a rich multipath environment, and more importantly, the propagation properties of this channel can be accurately modelled by a simple method based on theory of image. Computed by the measured 2×2 MIMO channel matrix,the CG very close to 2 can be obtained by this CCEMD antenna. Moreover, the measured MIMO performance is compared with the simulated result, which agrees well with each other.

In this paper, a CCEMD MIMO antenna based on a two-mode loop antenna having high isolation and pattern diversity is presented.

II. ANTENNA DESIGN AND RESULTS

2.1 Proposed antenna design

The proposed antenna is depicted in figure 1.A FR4 substrate with a dielectric constant of εr= 4.4 and loss tangent tanδ =0.02 is employed for its low cost. The radius and height of the substrate are Sr= 34 mm and hr= 0.8 mm, respectively. The proposed antenna consists mainly of two parts: the radiation loop based on segmented line with a circular patch as a ground on the right side and a ‘T’ shape microstrip feeding ports on the other side of the substrate. The segmented line is a kind of ZPSL, through which the current along the electric large antenna can be kept approximately unchanged in phase and flowing in a single direction. The reason we chose this ZPSL construction is to make the distance of two ports separate as far as possible to decrease the two ports coupling from physical structure. Here, the inner segment loop comprises eight arcs with width W1and angel α1,and the interspace arc angel is α2. With respect to the inner segment loop, the outer segment loop is rotated 22.5˚ with width W2. As shown in figure 1 (a), for port 1, a 50 Ω lump port is directly attached to the edges of the inner loop and a good input impedance matching and resonant frequency can be realized by adjusting W2and W1without any other matching constructions, a well input impendence about 49 - j1.9 Ω is obtained; for port 2, as can be seen in figure 1 (b), a microstrip line feeding construction which is composed of a T-shaped strip and a circular disk patch as the ground is used to feed the outer segment loop. This feeding construction can be seen as a kind of power divider [12], through which the current with the same magnitude and opposition at the end of line can be excited. The impedance matching and resonant frequency can be re-alized by adjusting the strip width Wp and length lp, respectively. The input impedance of port 2 is 50.8 + j0.1 Ω.

The antenna design and optimization are carried out by the commercial software High Frequency Structure Simulator (Ansoft HFSS).The design procedures are given as follows.

Step 1) Choose ZPSL construction for our purpose to decrease the mutual coupling of ports.

Step 2) Determine W1and circular patch ra-dius R1to tune the resonant frequency and impedance matching can be realized by adjusting W2.

Fig. 1. Geometry scheme of the proposed antenna. (a) . (b) .

Fig. 2. Simulated current distribution at 2.4 GHz. (a) Port 1 is fed and port 2 is matched. (b) Port 2 is fed and port 1 is matched.

Step 3) Determine Wpand lp of port 2 to achieve the resonant frequency and input impedance matching.

The detailed parameters of the proposed antenna are given in table 1.

2.2 Current distribution

The surface current distributions of the two ports are depicted in figure 2, the perimeter of the loop is approximately one wavelength at 2.4 GHz. As can be seen from figure 2 (a),when port 1 is fed and port 2 is matched, the currents along the loop are uniform and inphase, which is identical to that of a small loop antenna, thus, it can be considered as a magnetic dipole, which is vertical to the xy-plane and horizontally polarized. From figure 2 (b), when port 2 is fed and port 1 is matched, standing wave currents are distributed on the side arcs with current antinode at the port 1 and port 2 positions. Hence, it can be considered as the array of two arc dipole antennas with linearly polarized radiation in the x-direction. The current has less impact on port 1, so high isolation (lower than -40 dB in measurement) is achieved between port 1 and port 2.

2.3 Measurement and simulation results

A prototype of the proposed antenna was manufactured and its S-parameters were measured using Agilent vector network analyzer(E5071C). For port 1, a SMA connector with coaxial line is used to feed and a ferrite-bead is used to cover the feed cable near the port 1 as a balun during the measurement; for port 2,a SMA connector is utilized. According to the simulated and measured return loss of each port as shown in figure 3, it has simulated -10 dB impedance bandwidths of 300 MHz (2.27-2.57 GHz) at port 1 and 260 MHz (2.26-2.52 GHz) at port 2. The measured -10 dB impedance bandwidths are 420 MHz (2.32-2.76 GHz) at port 1 and 300 MHz (2.3-2.6 GHz)at port 2. There is a fairly good agreement between measurement and simulation results,and the small discrepancies are caused by the inevitable fabrication errors and measuring tolerance. The measured common -10 dB impedance bandwidth at the two ports is 280 MHz (2.32-2.6 GHz) to cover 2.4-2.48 GHz band and it is suitable for MIMO WLAN systems.

Figure 4 describes the simulated and measured results of the isolation between ports 1 and 2. The simulated mutual coupling is less than -58 dB over the whole working frequency band, however, the measured result is about-45 dB at the center frequency near 2.5 GHz.The discrepancies might be caused by the small inhomogeneity in dielectric constant of the manufactured substrate and effect of the SMA connector with a coaxial cable. Despite there are some differences between the simulated and measured isolation results, the measured isolation is still better than 41 dB over 2.4-2.48 GHz. The measured and simulated radiation patterns at the resonant frequency of 2.45 GHz are plotted in Figure 5. Both the xy-plane and yz-plane patterns are shown for ports 1 and 2, respectively, and good agreement between the simulated and measured results can be obtained. When port 1 is excited with port 2 connected to a 50-Ω match load,the antenna radiates a horizontal polarized omnidirectional pattern in the xy-plane and 8-shaped radiation pattern in the yz-plane,as shown in Figs. 5 (a) and (b). In contrast,when port 2 is excited with port 1 connected to a 50-Ω match load, the good horizontal polarized omnidirectional pattern in yz-plane and 8-shaped radiation pattern in the xy-plane are obtained, as shown in Figs. 5 (c) and (d).The measured cross-polarization values of the antenna are better than -15 dB, which are larger than the simulated values, maybe due to the coaxial cable-based feeding line and the non-soft test cable leading the test plane of the proposed antenna to be slightly unparalleled to the test polarization. Thus, the proposed antenna behaves like a CCEMD antenna and it is also a good candidate for MIMO system application with pattern diversity.

Fig. 3. Simulated and measured S-parameters of the proposed antenna.

Table II. Performance comparison for two-port MIMO antennas

Fig. 5. Measured and simulated normalized radiation patterns of the fabricated prototype at 2.45 GHz. (a) Port1, xy-plane. (b) Port1, yz-plane. (c) Port2, xy-plane.(d) Port2, yz-plane.

Fig. 6. Schematic diagram of the two parallel metal planes and the setup in measurement campaign.

The performance comparisons to other twoports MIMO antennas are given in table II.Compared with the antenna in [3] which has a 3D structure and the two antennas are assembled mechanically, the proposed antenna radiates with two different feed constructions and a common radiator which are fabricated on a PCB board with compact size. In addition, the proposed antenna with a different ZPSL structure has a wider impedance bandwidth that the common -10 dB impedance bandwidth at the two ports is 280 MHz (2.32-2.6 GHz) and only 120 MHz in [7], the isolation is lower than -41 dB and -25 dB in [7], respectively. It can be summarized that the proposed antenna has the advantages of simple co-located planer structure, high isolation, and better band coverage, which is suitable for MIMO system.

III. CHANNEL PERFORMANCE AND MODELING

The MIMO channel characteristic is measured for the proposed antenna installed at both the transmitting (Tx) and receiving (Rx) ends.The measurement experiment is conducted in the environment composed of two parallel metallic planes, as shown in figure 6, considering that this propagation is full of multipath reflections, and furthermore, the propagation response of the dual-polarized MIMO channel can be accurately modeled by the theory of image [13]. Two copper planes are used which have the same size of 1 m × 2 m (w × l ) with a vertical separation of h = 1.25 m. During the experiment, the plane containing the loop antenna is kept parallel to the yz-plane, as shown in figure 6. The complex transmission coeffi-cients between Tx and Rx antennas were measured by an Agilent vector network analyzer(E5071C).

The performance of a MIMO system is generally evaluated by the channel capacity,for a 2 × 2 MIMO system, it can be expressed as [14]

where I is a 2×2 identical matrix, SNR is the signal to noise ratio and H†means the transpose conjugate of H. Here, we assume a 20 dB SNR and a widely used channel normalization, where nTand nRdenote the number of Tx and Rx elements, respectively,denotes the Frobenius norm.

In the measurement setup, the ports not under test were terminated with 50-Ω match load. When the transmitting antenna is locat-ed at the point (0.25, 0.625, 0), the radiation modes of the proposed antenna are like a parallel magnetic dipole (blue arrow) and a vertical electric dipole (red arrow), as depicted in Figure 6. The receiving antenna is parallel to the Tx antenna with a communication distance R and same height. Here, Rx antenna is moved forward over a 2-m path with the same height of 0.625 m and 19 points were measured. By the measured channel matrix H, the MIMO channel capacity can be computed by (1),here, CG defined by the ratio between the 2×2 MIMO system over the corresponding SISO system is used as the metric of performance evaluation. The results of the CG computed by the measured channel matrix at 2.45GHz are illustrated in Figure 7, as a contrast, the CG values obtained by the theory of image in the channel containing two parallel perfect electronic conductor (PEC) planes are also computed for the CCEMD antenna composed of ideal electric and magnetic dipoles.

For a dipole in the channel containing two parallel PEC planes, located in y = hy, the image and real sources are regular located along axis y at the positions y = 2mh ± hy(m=0, ± 1, ± 2,...). The y polarized electric field component Eyand x polarized magnetic field component Hxat any point of this channel can be computed by the following expression

Wheredenotes the sum of the contributions to the total y electric field components produced by the sources located at the y+= 2mh + hy(m=0, ± 1, ± 2,...), the other summation terms in (2) have similar meanings to

Fig. 7. Comparison of CG between the experimental results and that from the theory of image.

In Figure 7, the simulated results of CG in free space (FS) channel are also given for comparison. Figure 7 shows that, when the separation distance R between the Tx and Rx antennas is less than 1 m (almost 8 λ at 2.45 GHz), the CG in FS channel slightly decrease with the increase of R, however, the values are just slightly larger than 1 and remain unchanged when R is larger than 1 m. Compared with the results in FS channel, the values of CG are obviously improved and very close to 2 in the channel containing two parallel PEC planes. The measured results of CG agree with the simulated results fairly well. It is obvious that two separate subchannels can be obtained using the proposed CCEMD MIMO antenna in a multipath rich environment.

IV. CONCLUSION

In this letter, a CCEMD MIMO antenna based on a two-mode loop antenna having high isolation and pattern diversity is presented. The radiation modes are like an electric dipole and a magnetic dipole that are co-located and co-polarized. The proposed antenna is fabricated and tested. Its measured common -10 dB impedance bandwidth at two the ports is 280 MHz (2.32 - 2.6 GHz) and the port isolation is better than 41 dB in this band. Meanwhile,its MIMO channel performance is measured in a multipath rich environment containing two parallel PEC planes, and the MIMO CG obtained by the CCEMD consisted of ideal electric and magnetic dipoles are also computed in FS and in the channel containing two parallel PEC planes by image theory. Results show that, by the CCEMD MIMO antenna,slightly larger than 1 time of capacity gain can be obtained over the SISO antenna in FS,however, in the multipath rich propagations,CG values very close to 2 can be obtained,which means that two independent subchannels can be achieved by using the proposed CCEMD antenna. The simulated results of CG by the theory of image are in good agreement with that from experimentation in the channel containing two parallel PEC planes.

ACKNOWLEDGMENTS

This work was supported by the National Natural Science Foundations of China(61771435).

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